Measuring parameters in radio frequency systems. Determining the signal power at the input of the receiver of the laboratory of the department How to measure the power of a radio signal of a certain frequency

SHORT DESCRIPTION

Power meters series Anritsu ML2490A They are high-speed digitizers and processors of signals coming from power sensors connected to them. The Anritsu ML2495A model is single-channel and supports the connection of one sensor, and the Anritsu ML2496A model can work simultaneously with two different sensors. Depending on the types of sensors connected, the frequency range can be from 100 kHz to 65 GHz.

Thanks very much high speed digitization (time resolution reaches 1 ns), Anritsu ML2490A series meters can be used to develop and configure radars, and the bandwidth of these devices, equal to 65 MHz, allows them to be used at all stages of construction and operation of 3G, 4G and 5G wireless communication systems, including next generation systems based complex technologies modulations such as OFDM.

In addition to pulse and peak power sensors, the Anritsu ML2490A series can connect a variety of sensors to measure stationary radio signals (CW), making them versatile in application. You can download a complete description of all the characteristics of the Anritsu ML2490A series below on this page in the section.

Main characteristics:
Number of channels: 1 (model ML2495A) or 2 (model ML2496A).
Frequency: 100 kHz – 65 GHz (depending on sensor).
Bandwidth (video band): 65 MHz.
Typical rise time: 8 ns (with MA2411B pulse encoder).
Time resolution: 1 ns. Built-in power calibrator (50 MHz and 1 GHz).
Ideal for radar applications and wireless networks(4G and 5G).
Power measurements: Average, Min, Max, Peak, Crest, PAE (Power Added Efficiency).
Screen 8.9 cm (resolution 320 x 240). Interfaces: Ethernet, IEEE-488 (GPIB), RS-232.
Weight: 3 kg. Dimensions: 213 x 88 x 390 mm. Operating temperature: from 0°C to +50°C.
Accurately measure the power of any radio signal

DETAILED DESCRIPTION

The Anritsu ML2490A series of RF power meters offers superior performance compared to Anritsu's other two meter series (ML2480B and ML2430A). The ML2490A series includes two models: the single-channel ML2495A and the dual-channel ML2496A. Both models work in conjunction with external sensors (sensors). Anritsu ML2490A power meters are compatible with six series of sensors that cover a very wide range of applications in the frequency range from 10 MHz to 50 GHz and in the power range from -70 dBm to +20 dBm.

Depending on the type of connected sensor, Anritsu ML2490A meters can measure the following signal strength parameters: Average (average value), Min (minimum value), Max ( maximum value), Peak (peak value), Crest (peak factor), Rise-time (rise time), PAE (Power Added Efficiency), etc. For calibrating sensors, Anritsu ML2490A devices are used as standard function contain a built-in power calibrator for two frequencies: 50 MHz and 1 GHz.

This photo shows the Anritsu ML2495A Single Channel RF Power Meter and the Anritsu ML2496A Dual Channel RF Power Meter along with two of the best sensors: the Anritsu MA2411 Pulse Sensor (up to 40 GHz) and the Anritsu MA2491A Wide Wide Sensor (up to 18 GHz).

Anritsu ML2495A single channel meter (top) and Anritsu ML2496A dual channel meter (bottom) along with the MA2411 pulse power sensor and MA2491A wideband power sensor.

Pulse power sensor Anritsu MA2411B

The Anritsu ML2495A and ML2496A power meters, together with the Anritsu MA2411B sensor, are ideal for measuring pulsed radio signals in the frequency range from 300 MHz to 40 GHz. With a typical rise time of 8 ns and a resolution of 1 ns, direct measurements of the characteristics of radar pulses, as well as a wide variety of other types of signals with a pulse or burst structure, are possible.

This photo shows a screenshot of the Anritsu ML2496A power meter with the results of measuring the parameters of the edge of an RF pulse. The measurements were carried out using an Anritsu MA2411B pulsed power sensor. The scale on the horizontal axis is 20 ns per division, and on the vertical axis 3 dB per division. The signal coming from the sensor was digitized at a speed of 62.5 MSa/s.

This photo shows a screenshot of the Anritsu ML2496A power meter showing the measurement results of four consecutive RF pulses. The scale on the horizontal axis is 2 µs per division, and on the vertical axis 5 dB per division. For each pulse, you can measure: rise time, fall time, duration and other parameters, including the pulse repetition interval PRI (Pulse Repetition Interval). The results for a group of pulses are also displayed on the screen: minimum, maximum and average power values.

Measurement of parameters of four successive radio frequency pulses.

When measuring high-power radio signals, attenuators or couplers are often used. The Anritsu ML2490A series power meters have the ability to automatically take into account the value of an external attenuator or coupler so that the measurement results on the screen correspond to the actual power.

Before using the Anritsu MA2411B sensor with the ML2490A series power meter, they must be calibrated together. To do this, a reference signal output (Calibrator) with a frequency of 1 GHz and an amplitude of 0 dBm (1 mW) is located on the front panel of the power meter. By connecting the sensor to this output and clicking the corresponding menu item, you will calibrate the sensor and zero the errors of the measuring path, which will prepare the device for accurate measurements.

The Anritsu MA2411B sensor is optimized for measuring pulse signals and wideband modulated signals, but can be successfully used to accurately measure the characteristics of stationary (CW) and slowly varying radio signals. The corresponding screenshot is shown in this photo.

Wideband power sensors Anritsu MA2490A and MA2491A

Two wideband sensors are designed to measure the parameters of telecommunication signals, as well as some types of pulse signals: Anritsu MA2490A (from 50 MHz to 8 GHz) and Anritsu MA2491A (from 50 MHz to 18 GHz). Both sensors provide 20 MHz bandwidth (also called video bandwidth or response rate), which is sufficient to accurately measure rapidly changing signals such as 3G/4G, WLAN, WiMAX and pulses from most types of radar systems. The rise time for these sensors in pulsed measurement mode is 18 ns.

Impulse characteristics The MA2490A and MA2491A sensors are slightly worse than the MA2411B discussed above, but the minimum measured power is -60 dBm, instead of -20 dBm for the MA2411B. A significant expansion of the lower power threshold is achieved due to the presence of an additional measuring path inside the sensors, which is automatically turned on at low power values.

This photo shows a screenshot of the Anritsu ML2496A power meter with the results of measuring GSM signal parameters. Measurements were carried out using an Anritsu MA2491A wideband power sensor. The scale on the horizontal axis is 48 µs per division, and on the vertical axis 5 dB per division. The peak power of individual signal fragments reaches 12 dBm.

Measuring GSM signal parameters using the Anritsu MA2491A wideband sensor.

High-precision diode power sensors (sensors) of the Anritsu MA2440D series

This series of high-precision sensors is designed for radio signals with a low rate of change or modulation (such as TDMA), as well as stationary (CW - Continuous Wave) signals. The response speed (video bandwidth) of these sensors is 100 kHz and the rise time is 4 µs. All MA2440D series sensors have a built-in 3 dB attenuator, which significantly improves the matching (SWR) of the sensor's RF input connector. A wide dynamic range of 87 dB and linearity better than 1.8% (up to 18 GHz) and 2.5% (up to 40 GHz) make these sensors ideal for a wide range of applications, including radio gain and attenuation measurements.

The Anritsu MA2440D series of sensors consists of three models, differing in the upper frequency range and type of input connector: model MA2442D (from 10 MHz to 18 GHz, N(m) connector), model MA2444D (10 MHz to 40 GHz, K(m) connector) and model MA2445D (10 MHz to 50 GHz, connector V(m)). As an example, this photo shows an Anritsu MA2444D sensor with a K(m) connector.

High-precision power sensors based on thermal effect of the Anritsu MA24000A series

This series of high-precision sensors is designed for stationary (CW - Continuous Wave) and slowly changing radio signals. The rise time for these sensors is 15 ms. The operating principle of sensors in this series is based on the thermoelectric effect, which allows you to accurately measure the average power of any radio signal, regardless of its structure or type of modulation. Dynamic range of these sensors is 50 dB, and linearity is better than 1.8% (up to 18 GHz) and 2.5% (up to 50 GHz).

The Anritsu MA24000A series of sensors consists of three models, differing in the upper frequency range and type of input connector: model MA24002A (from 10 MHz to 18 GHz, N(m) connector), model MA24004A (10 MHz to 40 GHz, K(m) connector) and model MA24005A (10 MHz to 50 GHz, connector V(m)). All three Anritsu MA24000A series sensors are shown in this photo.

Operating principle and internal structure of Anritsu ML2490A series power meters

Power sensors connected to the Anritsu ML2490A series meters perform the function of converting the high-frequency signal, the power of which must be measured, into a low-frequency signal. This low-frequency signal comes from the sensor to the input of the ML2490A series meter, is digitized using the built-in ADC, processed by a digital signal processor and displayed on the device's display.

This figure shows the block diagram of the single-channel ML2495A. In this block diagram, two ADCs (analog-to-digital converters) are highlighted in green, with the help of which the low-frequency signal coming from a power sensor connected to the meter is digitized. If a diode sensor of the Anritsu MA2440D series or a thermoelectric sensor of the Anritsu MA24000A series is connected, then digitization is performed using a 16-bit ADC. And if an Anritsu MA2411B pulse sensor or Anritsu MA2490A or MA2491A wideband sensors are connected, then digitization is performed using a high-speed 14-bit ADC.

Structural scheme single-channel power meter Anritsu ML2495A.

And this is what the internal structure of the Anritsu ML2490A series power meter looks like. In the center there is a small rectangular board of a built-in calibrator for 50 MHz and 1 GHz, the high-frequency cable from which is connected to the N connector on the front panel. Under the calibrator board there is a large measurement board containing an analog part, an ADC and an array of programmable logical matrices. Immediately below the measurement board there is a second large digital processing and control board containing a DSP (digital signal processor), a microcontroller and digital display and control units.

All Anritsu ML2490A series power meters come complete with computer program remote control Anritsu PowerMax. This program runs on Windows compatible personal computer and allows you to remotely control the operation of a single-channel Anritsu ML2495A or dual-channel Anritsu ML2496A device. Taking measurements with PowerMax software makes it easy initial setup device, speeds up measurement processing and allows convenient documentation and storage of results.

An example of the Anritsu PowerMax main window is shown in this screenshot. In this case, a two-channel Anritsu ML2496A model is controlled, the first channel of which is connected to an Anritsu MA2411B pulse power sensor, and the Anritsu MA2491A broadband power sensor is connected to the second channel. To enlarge the image, click on the photo.

Anritsu ML2490A series power meters come with Anritsu PowerMax software.
Click on the photo to enlarge the image.

Specifications of Anritsu ML2490A meters and power sensors

Below is a list of the main technical characteristics of the Anritsu ML2490A series power meters. For detailed technical characteristics of the meters, see below on this page in the section.

Main technical characteristics of Anritsu ML2490A series power meters.

Below is a list of the main technical characteristics of power sensors (power sensors) various types, which are compatible with Anritsu ML2490A series meters. For detailed technical characteristics of the sensors, see the section below on this page.

Main characteristics of power sensors compatible with Anritsu ML2490A series.

Anritsu ML2490A Series Power Meter Package Contents

Options and Accessories for Anritsu ML2490A Series Power Meters

Main options:
- option 760-209 (hard transport case for transporting the device and accessories).
- option D41310(soft bag for transporting the device with a shoulder strap).
- option 2400-82 (rack mounting kit for one meter).
- option 2400-83 (rack mount kit for two meters).
- option 2000-1535 (protective cover for front panel).
- option 2000-1536-R(0.3 meter cable for connecting the measuring sensor).
- option 2000-1537-R(1.5 meter cable for connecting the measuring sensor).
- option 2000-1544 (RS-232 cable for flashing the device).

Compatible power sensors:
- sensor Anritsu MA2411B(pulse sensor from 300 MHz to 40 GHz, from -20 dBm to +20 dBm).
- sensor Anritsu MA2490A(wideband sensor from 50 MHz to 8 GHz, from -60 dBm to +20 dBm).
- sensor Anritsu MA2491A(wideband sensor from 50 MHz to 18 GHz, from -60 dBm to +20 dBm).
- sensor Anritsu MA2472D(standard diode sensor from 10 MHz to 18 GHz, from -70 dBm to +20 dBm).
- sensor Anritsu MA2473D(standard diode sensor from 10 MHz to 32 GHz, from -70 dBm to +20 dBm).
- sensor Anritsu MA2474D(standard diode sensor from 10 MHz to 40 GHz, from -70 dBm to +20 dBm).
- sensor Anritsu MA2475D(standard diode sensor from 10 MHz to 50 GHz, from -70 dBm to +20 dBm).
- sensor Anritsu MA2442D(high precision diode sensor from 10 MHz to 18 GHz, from -67 dBm to +20 dBm).
- sensor Anritsu MA2444D(high precision diode sensor from 10 MHz to 40 GHz, from -67 dBm to +20 dBm).
- sensor Anritsu MA2445D(high precision diode sensor from 10 MHz to 50 GHz, from -67 dBm to +20 dBm).
- sensor Anritsu MA2481D(universal sensor from 10 MHz to 6 GHz, from -60 dBm to +20 dBm).
- sensor Anritsu MA2482D(universal sensor from 10 MHz to 18 GHz, from -60 dBm to +20 dBm).
- sensor Anritsu MA24002A(thermoelectric sensor from 10 MHz to 18 GHz, from -30 dBm to +20 dBm).
- sensor Anritsu MA24004A(thermoelectric sensor from 10 MHz to 40 GHz, from -30 dBm to +20 dBm).
- sensor Anritsu MA24005A(thermoelectric sensor from 10 MHz to 50 GHz, from -30 dBm to +20 dBm).

Documentation

This documentation is in PDF format contains the most Full description capabilities of Anritsu ML2490A series power meters, their technical characteristics and operating modes:

Description of Anritsu ML2490A power meters and sensors for them (in English) (12 pages; 7 MB)

Technical characteristics of Anritsu ML2490A meters and sensors for them (in English) (12 pages; 1 MB)

Anritsu ML2490A Power Meters User Manual (English) (224 pages; 3 MB)

Anritsu ML2490A Meter Programming Guide (English) (278 pages; 3 MB)

Brief information about devices for measuring the power of radio signals (in English) (4 pages; 2 MB)

And here you can find our tips and other useful information on this topic:

Brief overview of all Anritsu RF test instrument series

Brief overview of all Anritsu portable RF analyzer series

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Purpose: to study the instrument arsenal of the department's laboratories and the main factors that determine the energy of radio lines.

Satellite communication and broadcasting lines consist of two sections: a transmitting earth station (ES) - a relay on an artificial Earth satellite (AES) and an AES relay - a receiving ES. The signal power at the input of the ES receiver can be determined from the formula that is used to calculate any line-of-sight radio links:

Where P prd– power at the output of the satellite repeater transmitter,

γ prd and γ prm– transmission coefficients of the paths connecting, respectively, the output of the transmitter with the transmitting antenna on the satellite and the output of the receiving antenna with the receiver of the satellite,

G prd And G prm- the gains of the transmitting and receiving antennas, respectively,

L o And L extra– main and additional losses of signal energy in the space between the satellite and the station.

Major losses L o caused by energy dissipation in free space when moving away from the emitter

, (2.2)

where λ is the electromagnetic wavelength

, (2.3)

f– frequency of the transmitter signal, c ≈ 3∙10 8 m/sec – speed of propagation of electromagnetic waves,

d– distance between satellite and station.

Distance d between satellite and station depends on altitude H satellite orbit, which determines the size of the satellite's visibility zone.

The satellite visibility zone is the part of the Earth's surface from which the satellite is visible during a given duration of a communication session at an elevation angle of at least a certain specified angle.
.

The instantaneous visibility zone of an artificial satellite is the visibility zone at a certain point in time, i.e. with zero communication session duration. When an satellite moves, the instantaneous visibility zone moves, so the visibility zone during a communication session is always less than the instantaneous one. The size of the instantaneous visibility zone can be estimated by the length of the arc
or corners And (Fig. 2.1).

Corner represents the angular distance of the zone boundary from the sub-satellite point (relative to the center of the Earth), and the angle equal to half the maximum angular size of the visibility zone relative to the satellite located at the point . Points And are on the border of the visibility zone and are distant from the satellite at a distance
, called the maximum slant range.

For a triangle ∆
the following relations are valid:

, (2.4)

, (2.5)

Where R Z=6400 km – radius of the Earth.

Additional losses L extra caused by the atmosphere, precipitation and other reasons.

Antenna gains when using parabolic mirror antennas with a mirror diameter D determined from the expression:

. (2.6)

Task 2. Using formulas (2.1) – (2.6) determine the signal power at the input of the receiver of the station located on the border of the visibility zone. The initial data for the calculation are given in Table 2.1. The assignment option is determined by the teacher.

Table 2.1

Using expressions (2.4) – (2.5) determine the distance d between satellites and satellites.

Substitute the necessary data into expression (2.1).

Task 3. Determine the signal power at the input of the satellite receiver located at the sub-satellite point S (Fig. 2.1). The initial data and calculation procedure are the same as for task 2.

Compare the results obtained in task 2 and task 3.

Report must contain the characteristics and description of the department's antennas, as well as the results of calculations for tasks 1-3.

WORK IN THE COMPUTER LABORATORY

SIMULATION

The goal of the students’ work is to acquire programming skills in the MatLab environment.

To enter the MatLab environment, move the mouse pointer to the software system logo and double-click the left mouse button (LMB).

Exercise. Construction of a Simulink model of the stand.

The transition to the Simulink package can be done in two ways:

after entering the MatLab environment in command line in the control window opposite the pointer, the simulink command is typed;

using the mouse - one left-click on the blue-red-black symbol containing an arrow.

After these actions, the library window (Library:Simulink) and the yet untitled field window will open, on which the model will be assembled. In the seventh version of MatLab, to create such a field after entering Simulink, you need to click LMB on the blank sheet symbol.

First, students should become familiar with the sections of the Simulink library: Sources - sources; Sinks – loads, and also independently find sections containing blocks Abs, F cn, Relational Operator, Mux, etc.

The blocks necessary for assembling the block diagram are dragged with the mouse from the library sections while pressing the LMB.

Models of assembled stands are shown in Fig. 3.1. Figure 3.1a shows a model containing two harmonic signal shapers. The sine function argument forms the Ramp block.

To set the parameters of this and other blocks, first select the block by clicking LMB, and then double-clicking to open a window into which the corresponding parameters are entered. The Slope parameter of the Ramp source is set to pi /50 (in MatLab language the constant
written as pi).

By using the Mux block, the Scope oscilloscope becomes dual beam. Students choose the parameters of oscilloscope models independently. Set the simulation time (Stop time) to 100: Simulation – LMB click, Parameters – LMB click, record the time in the Stop time column.

The program can also be launched using the mouse: Simulation – left-click, Start – left-click. You can also run the program by clicking on the triangle icon.

It is necessary to sketch (print) block diagrams of models and observed oscillograms.

Figure 3.1b shows a model of a comparator - a device that generates a single signal when the condition specified on the comparison device block - Relational Operator - is met.

By selecting the assembled model and using the Create Subsystem command in Edit mode, you can make the comparator model a Subsystem block. Such a block is shown in Fig. 3.1c, which shows a model of a device for comparing signal levels of Sine Wave and Constant sources. In this simulation experiment, the amplitude of the harmonic vibration is 1, the angular frequency is 0.1
with simulation time – 100.

Draw (print) a diagram of the model and oscillograms.

Individual tasks are given in Table 3.1. The structural diagram of the models for all options is the same. It is obtained from the block diagram shown in Fig. 3.1a, if the Fcn 2 block and the Mux block are excluded from the latter. Thus, the output of the Ramp block is connected to the input of the Fcn 1 block, and the input

Scope oscilloscope is connected to the output of block Fcn 1.

The simulation time for all options is 100.

Report for this section must contain:

block diagrams of the studied Simulink models;

oscillograms;

Table 3.1

7.9.Measurement of parameters in radio frequency systems BER function measurement (C/N)

In modern measurement technique BER uses various schemes, of which two main ones can be distinguished.

Rice. 7.16. Scheme of the tunable attenuator method.

In this method, a tunable attenuator is included in the radio frequency path of the receiver, with the help of which additional attenuation is introduced, and the stability of the reception signal is assumed to be constant throughout the entire measurement time. The signal and noise levels are measured using a power meter, while measuring the noise in the intermediate frequency path of the receiver without filtering gives a value greater than the actual noise power in the operating band of the path. Therefore, when measuring power, additional filters tuned to the operating frequency band are used.

The BER error parameter is measured by a digital channel analyzer.

The main disadvantage of the method is the assumption of constant power useful signal during the entire measurement period. In real conditions, the level of the useful signal undergoes significant fluctuations due to multipath propagation of radio waves and changes in propagation conditions. For this reason, the C/N ratio can also change, and even a 1 dB change in C/N can cause a change in BER by an order of magnitude. Thus, this method does not provide the required measurement accuracy, especially for low BER values.

2. Interference method for measuring BER(C/AT), the diagram of which is shown in Fig. 7.17, uses a special device - an analyzer/simulator of the C/N parameter, which measures the power level of the useful signal C when introducing a given noise level N, which ensures high accuracy in determining the C/N parameter. IN this method the analyzer/simulator automatically adjusts the level of introduced noise, and the measurement accuracy of the BER(C/AT) characteristic can reach values ​​of ~1СГ12. In conclusion of this consideration of the BER (CIN) function, we note the following.

1. Comparison of theoretical and practical dependences VESHCHS/N) show that practical dependences differ from theoretical ones in that for practical BER values ​​a larger C/N ratio is required. This is due to various reasons for parameter degradation in the intermediate and radio frequency paths.

2. In practice, the contributions of the radio and intermediate frequency paths are comparable to each other, while for transmission systems digital information at speeds up to 90 Mbit/s, the following BER degradation levels are observed.

Rice. 7.17. Scheme of the interference method for measuring BER(C/N)

Deterioration in the IF intermediate frequency path:

Errors in phase and amplitude of the modulator - OD dB;

Intersymbol interference due to filter operation - 1.0 dB;

Presence of phase noise - 0.1 dB;

Differential encoding/decoding procedures - 0.3 dB;

Jitter (phase jitter) - 0.1 dB;

Excess noise bandwidth of the demodulator - 0.5 dB;

Other reasons (aging effect, temperature instability) - 0.4 dB.

So, in total, the deterioration in BER in the IF path can reach 2.5 dB. BER degradation in the radio frequency path:

Nonlinearity effects - 1.5 dB;

Impairments due to channel bandwidth limitation and group delay time - 0.3 dB;

Interference in adjacent channels - 1.0 dB;

Impairment due to attenuation and echo effects - 0.2 dB. In total, in the RF radio frequency path the BER degradation will be 3 dB, that is, the total in the system

Transmission BER degradation can reach -5.5 dB.

It should be noted that in the diagrams of Fig. 7.16, 7.17 the purpose of equalizers in digital radio paths was not considered.

Frequency and power measurements in radio frequency paths.

Measurements of the frequency and power of a useful radio signal are implemented in practice using the following methods:

1)frequency meters and power meters are used,

2) spectrum analyzers with marker measurement functions are used.

In the second method, the marker provides movement along the spectral characteristic while simultaneously displaying the values ​​of the frequency and power parameters of the useful radio signal.

To expand the capabilities of measuring power parameters, modern spectrum analyzers provide spectral smoothing, noise filtering, etc.

Analysis of the operation of equalizers.

Compared with cable systems radio air, as a medium for transmitting radio signals, has characteristics that randomly change over time. Due to the widespread use of digital radio communication systems and increased requirements for the accuracy of their transmission, equalizers are included in receiving devices to dramatically reduce the influence of multipath propagation (signal alignment) and group delay time (signal auto-tuning). When using digital methods for modulating high-frequency signals, developers encountered difficulties in accurately tuning modems and other channel-forming devices as part of the radio frequency path. In this case, equalizers also act as elements of compensation for possible nonlinearities in devices of the radio frequency transmission path. In modern radio frequency information transmission systems, there are two main types of attenuation associated with factors of radio signal propagation along the radio frequency path.

1) Linear attenuation, which is a frequency-independent uniform decrease in the signal amplitude from signal distribution factors. Linear attenuation is usually caused by natural factors in the propagation of electromagnetic waves:

With through distribution in forest areas;

When distributed in the atmosphere in the presence of hydrometeors (rain, snow).

2) Attenuation due to multipath propagation of radio signals.

These two factors change the amplitude of the desired signal, leading to a change in the C/N ratio, which ultimately affects the BER error parameter. Changes in the structure of the useful signal associated with these two attenuations are compensated by equalizers. As you know, the basis of the operation of any equalizer is the use of a narrow-band notch filter to eliminate the nonlinearity of the useful signal. The main measurement parameter is the dependence of the filtering depth on frequency at given parameter BER, called the M curve or W curve in various reviews (Fig. 7.18).

Rice. 7.18. M curves for cases of absence and presence of an equalizer.

To obtain the M curve, various signal transmission conditions are usually simulated, which are compensated by an equalizer and in the process of compensation, the M curve is constructed. The measurement scheme is shown in Fig. 7.19.

As a result of the measurements, diagrams are obtained in the form of two-sided curves M, of which one is hysteresis-free (showing the ability of the equalizer filter to provide a filtering depth at a given frequency sufficient to level the structure of the useful signal) and the other is hysteresis (showing the performance of the filter during its actual operation, if necessary first increasing and then decreasing the filtration depth parameter). In practice, both types of curves are essential for analyzing equalizer performance.

Rice. 7.19. Measuring scheme for M curves

Measurements of parameters of phase-frequency characteristic unevenness and group delay time.

The unevenness of the phase-frequency response (PFC) of the radio frequency path is determined by the group delay time (GDT) from the formula:

Direct measurement of the dependence of the phase shift on frequency f(n) and subsequent differentiation of the resulting dependence is implemented, as a rule, for systems with a low level of phase noise; however, for radio communication systems, phase noise is present in the channel, which leads to uneven phase response and a change in the group delay. Typically, group delay measurements are carried out during acceptance tests of radio systems and take into account possible deviations in the operation of the transmitter, receiver, antenna devices and radio signal propagation conditions. The paper describes two methods for measuring group delays based on the use of composite radio signals.

Measurements of immunity to linear fading and multipath attenuation of radio signals

The parameters of radio signals change due to linear attenuation and attenuation caused by multipath propagation of radio signals. When carrying out factory tests, an acceptable limit of linear attenuation is introduced, not exceeding 50 dB for BER = 10~3. To compensate for linear attenuation, equalizers are used as part of the transmitter/receiver. The performance of an equalizer that compensates for linear attenuation can be measured using tunable attenuators.

When measuring resistance to attenuation associated with multipath propagation of radio signals, it is possible to use a state diagram and an eye diagram that display:

State diagram - crosstalk between I and Q signals is shown as ellipses,

Eye diagram - the phenomenon of multipath is reflected by the displacement of the centers of the “eyes” from the center to the edges.

However, both the state diagram and the eye diagram do not provide all the necessary measurement specification. To carry out practical measurements of the effectiveness of compensation for the phenomenon of multipath signals, methods are used that are consistent with the compensation methods. Since it is almost impossible to predict the occurrence of the multipath factor, the impact of this factor is taken into account using stress methods, that is, by simulating the phenomenon of multipath signal propagation. As noted in the work, two models for simulating multipath signal propagation are used.

1.Double-beam model. The modeling principle comes down to the theoretically based assumption that the attenuation is associated with two-beam interference, and the interfering beam has a delay (for the reflected beam) in time. From the characteristics of the unevenness of the frequency response (amplitude-frequency characteristic) and group delay for two-beam propagation of a radio signal it follows:

Reducing amplitude with changing frequency;

Changes in group delay and frequency response in the case of a minimum phase (when the main radio beam has a large amplitude);

Changes in the frequency response and group delay in the case of a non-minimum phase (when the resulting beam after the interference of two beams exceeds the main signal in amplitude).

2. Three-beam model. Since the two-beam model does not describe the phenomenon amplitude modulation and the appearance of weak beat patterns within the working frequency range, as a result of which the amplitude of the useful signal deviates within the operating range even if the beat node is outside the operating range, then a three-beam model is used to take into account the effect of amplitude shift. Typically, the two-beam model is used for high-quality measurements, and the three-beam model is used for precise measurements.

Intermodulation interference analysis.

When radio signals propagate in a path, intermodulation interactions of signals arise during multiplexing and demultiplexing, as well as under the influence of nonlinearities of channel-forming devices within the path. Typically, intermodulation distortion is quite low level- less than 40 dB relative to the level of the useful signal. However, controlling intermodulation distortion and eliminating its causes provides, in some cases, a solution to the problem of interference in adjacent channels. Spectrum analyzers are used to analyze intermodulation.

Measurements of characteristics of channel-forming radio frequency paths.

In addition to complex measurements, measurements of the characteristics of channel-forming radio frequency paths are widely used in practice, knowledge of which is necessary when designing and operating radio engineering information transmission systems. In addition to frequency and power measurements in the service area, there is a need to measure antenna systems, thermal noise levels, frequency stability of master oscillators, phase jitter, parameters of modems and amplification paths along with filtering devices.

Antenna system measurements.

Antenna-feeder devices as part of the radio frequency path play an extremely important role. The main parameters: radiation power, radiation pattern in the corresponding planes, gain, impedance, etc., are usually calculated and measured at the antenna production stage. During operation, important parameters are

Traveling wave coefficient (TWC): CBW = Umin/Umax, (7.38)

Standing wave ratio (SWR): SWR = 1/KBW, (7.39)

Level return losses from the antenna input, where Umin and Umax are the minimum and maximum voltages in the feeder line.

In the case of ideal path matching: transmitter output - feeder - antenna input, KBV = 1 (since all the energy from the transmitter output is directed to the antenna and at the same time £/min = Umax), in the case of Umin = 0, VSWR = oo KBV = 0 — a standing wave mode occurs in the feeder, which is unacceptable.

In a real case, SWR can take values ​​of 1.1...2, that is, SWR = 0.5...0.9. In radio paths of digital information transmission systems with digital types of modulation, a low level of return losses is required, that is, a minimum SWR value of -1.1, when the mode in the feeder line is close to a high degree of matching.

For example, for microwave links using 64 QAM modulation, the recommended antenna return loss suppression level is 25 dB or higher. To measure return losses, the circuit shown in Fig. is usually used. 7.20.

A signal is supplied from the microwave oscillator to the antenna through a passive directional coupler. In the presence of a wave reflected from the input, electromagnetic oscillations enter a spectrum analyzer (or selective receiver) through a directional coupler, where the level of reflected power is measured. To reduce the level of reflected power, the antenna-feeder path is matched. When used in practice instead of a power meter spectrum analyzer, the measurement accuracy decreases, since, together with the reflected signal, the power meter takes into account the level of noise associated with external influences on the radio channel in a given operating frequency range.

Measurements of the level of intrinsic thermal noise of radio frequency path elements.

As the noise level increases, intersymbol distortion increases sharply digital signals and the BER value increases. In state diagrams and eye diagrams, this is reflected in the increase in the size of the state display points and the effect of “closing the eyes.” Noise measurement various devices as part of the radio frequency path, it is performed at the operational stage to localize the point of increased noise level. Considering that the intrinsic noise of various devices in the radio frequency path is small, differential methods are used for measurements. To do this, an interfering single-frequency signal is mixed into the test signal and then noise measurements are made by the difference between the interfering signal and noise. This method is used when measuring low power noise. As an example in Fig. Figure 7.21 shows the results of noise measurements against the background of an interfering single-frequency signal for 16 QAM modulation at a signal-to-noise ratio C/I = 15 dB, while, as can be seen from the figure, an increase in the noise level leads to an increase in the size of the points on the state diagram and the effect of “closing the eye” " on the eye diagram.

Rice. 7.21. Examples of state diagram and eye diagram when measuring noise at C/1 = 15 dB.

Phase jitter measurements.

An important parameter for measuring radio frequency transmission systems with digital modulation is the phase jitter of the signal from the master oscillators of the receiver/transmitter, the so-called jitter. To analyze jitter, a state diagram is effectively used, since the eye diagram is not sensitive to it. If phase jitter of the signal occurs in the path, then, as follows from

Rice. 7.22, there is an increase in the size of the points of the state diagram. To eliminate problems associated with the presence of jitter when measuring jitter, additional measurements of the operating parameters of the master oscillators are usually performed and faults are eliminated.

Modem parameters measurements.

To measure modem parameters, analyzers are usually used that provide signal measurements in the form of state diagrams and eye diagrams, which give the most full information about the structure and changes in digital modulation parameters. In Fig. Figure 7.23 shows as an example a state diagram and an eye diagram for the case of quadrature amplitude modulation with 16 states 16 QAM, from which it follows:

Blurring of the state diagram points indicates the influence of noise;

Distortion in the size of the “eye” indicates possible disturbances in the operation of the digital channel (for example, the occurrence of intersymbol distortions).

Rice. 7.23. Example state diagram and eye diagram for 16-state AM 16 QAM case

Let's consider the following types of modem malfunctions and the corresponding diagrams.

1. Loss of synchronization in the digital channel.

Global demodulator failure/disconnection or phase locking failure can lead to a loss of matching between the modulator and demodulator and loss of signal in the transmission system. In this case, the state diagram is random distribution signals at the corresponding modulation levels, the “eye” of the eye diagram is completely closed (Fig. 7.24).

Rice. 7.24. An example of loss of synchronization in a digital channel: the state diagram represents a random distribution of signals into the corresponding modulation levels, the “eye” of the eye diagram is completely closed.

2. Violation of the settings of the modulation/demodulation level parameters.

In Fig. Figure 7.25 shows a state diagram, from which it follows that when the modulation/demodulation levels were established, an imbalance in the signal amplitude arose. Changes in the state diagram may indicate nonlinearities in the modulator or a malfunction of the DAC.

Rice. 7.25. An example of a violation of the modulation/demodulation level settings.

3. Violation of the orthogonality of the I and Q vectors of the demodulator.

One of the common malfunctions in the operation of the modem is a malfunction of the demodulator when the vectors I and Q polar coordinates demodulator are not strictly orthogonal. This leads to a discrepancy between the states and the orthogonal coordinate grid on the state diagram (Fig. 7.26).

This fault may or may not be accompanied by a phase synchronization error in the carrier recovery circuit. In the absence of an error, the result of the impact of this malfunction on the eye diagram is reduced to the closing of the “eye” on the diagram on signal I and the absence of any change on the Q diagram. If there is an error, the “eyes” of both diagrams will be closed. It should be noted that analysis of the eye diagram alone does not allow us to determine the cause of the malfunction, since this diagram completely coincides with the eye diagram in the presence of a high level of additive noise in the channel. In this case, only a state diagram can provide a reliable determination of the cause of the malfunction. Eliminating the described malfunction requires adjusting the demodulator in terms of the orthogonality of the I and Q signals. In the state diagram in Fig. 7.27 noted the presence of a phase synchronization error of 2.3 degrees.

Rice. 7.27. An example of a phase synchronization error occurring.

Measurements of operating parameters of amplifiers as part of the radio frequency path.

The main measured parameters of the operation of amplifiers as part of the radio frequency path are:

Noise introduced by amplifiers;

Nonlinearity parameters of amplification sections.

Amplitude overload can lead to the amplifier transitioning to a nonlinear mode and, as a result, a sharp increase in the probability of error in digital system transfers. The use of state diagrams and eye diagrams makes it possible to evaluate the reasons for the decrease in radio communication quality parameters (nonlinear distortions lead to blurring of the points of the state diagram and the closing of the “eye” of the eye diagram).

The main parameter of a radio transmitting device is the power of the signal emitted into the air. It should be noted that the requirements for signal power in the VHF range are dictated by the characteristics of radio wave propagation in this frequency range.

The first feature of the VHF range is the rectilinear propagation of radio waves within line of sight. Figure 1 illustrates this feature of radio wave propagation in this range.

Approximately, taking into account the refraction of radio waves in the VHF range, the line of sight range in kilometers L is determined as:

At antenna elevation height base station and repeater 70 m, communication range cannot exceed 70 km:

When the height of the base station antenna and repeater is 70 m, the communication range cannot exceed 70 km. Approximate line of sight ranges in the VHF range are shown in Figure 2.

Let's calculate the required for a given distance output power transmitter signal. To do this, we will use the well-known formula for determining the signal power at the input of a radio receiver:

prm is the gain of the radio receiver antenna (in times);

It should be noted that in mobile communication systems, signal strength is measured in dBm. This is the ratio of the absolute value of the signal power, expressed in watts, to the signal power of 1 mW.

Let us express from it the power required from the transmitter when operating in free space. For the 160 MHz band and omnidirectional antennas, this power will be equal to:

With a signal-to-noise ratio at the demodulator input of 6 dB, the transmitter power can be limited to 1 mW.

On the other hand, when a radio wave propagates along the surface of the earth, it experiences additional absorption. To explain the phenomenon of radio waves bending around various obstacles and their penetration into the shadow and penumbra regions, the Huygens-Fresnel principle is used. In accordance with the Fresnel model, the range of propagation of radio waves between the transmitting and receiving devices is limited by an ellipsoid of rotation around the line connecting them. This ellipsoid is multilayered and can include an infinite number of zones.

The zone closest to the line connecting the transmitter to the receiver is called the first Fresnel zone. It is generally accepted that during the propagation of radio waves, the most significant is the first Fresnel zone. About half of the transmitted energy is concentrated in it. Figure 3 shows a longitudinal section of the first Fresnel zone.

For any point on the radio link, the radius of the first Fresnel zone (R0) can be found using the formula:

When taking into account the influence of the Earth's surface, the largest radius of the first Fresnel zone is important. With the same height of the antennas, this radius will be in the middle of the radio link. In this case, formula (6) is transformed to the following form:

When the radio link range is more than 5 km, it is necessary to additionally take into account the curvature of the Earth as an obstacle. This effect is illustrated in Figure 3. To take into account the increase in the level of the earth's surface in the middle of the radio link due to its curvature, you can use the following formula:

The height values ​​of the obstacle created due to the curvature of the Earth for relative distances r tek /L are given in Table 1.

Table 1

Now let's calculate the additional absorption of the signal due to its shading by the Earth's surface. To do this, we calculate the height h max in the center of the radio path:

The relative clearance of the radio line will be equal to

Now, using the graph of the dependence of the signal attenuation relative to the clearance of the obstacle shown in Figure 4, we will determine the additional signal attenuation.

For a relative radio link clearance of -0.37, the additional signal attenuation will be 50 dB. As a result, the required transmitter power increases from -6 dBm to +44 dBm. This power corresponds to a transmitter power of 20 W.

In this case, we considered a situation where a single radio transmitter is located in one place. However, there are not many places convenient for placing base station repeaters. Therefore, usually a large number of radio transmitters of radio systems for various purposes are concentrated in one place. To ensure that they do not interfere with each other, various decoupling devices, such as filters, circulators, and combiners, have to be installed at the transmitter output. Each of them weakens the power of the radio signal. In addition, the signal can be attenuated by the antenna-feeder path. The total signal attenuation value can reach 12 dB. This leads to the fact that even if the power at the transmitter output is 100 W, then only 6 W will reach the antenna:

For illustration, let's convert this value to watts:

conclusions

• To operate in the VHF range, taking into account the influence of the curvature of the earth's surface and obstacles, a transmitter power of at least 2 W is required
• For stationary radio stations, the required power increases to 50 ... 100 W due to losses in feeders and combiners

Literature:

Other parameters of radio transmitting devices:

A very important characteristic of a radio transmitting device is the range of emitted frequencies. To organize mobile radio communications in the VHF range...
http://site/UGFSvSPS/DiapPrdFr/

When forming a radio signal, it is very important that the entire spectrum of the emitted signal is concentrated within the frequency band allocated for a given radio channel...
http://site/UGFSvSPS/maska/

Exercise. 3

Theoretical part. 4

Basic provisions. 4

Units for measuring radio signal levels. 5

Okamura-Hata model. 7

Model COST231-Hata. 8

Model COST 231-Walfisch-Ikegami. 8

Research results. eleven

Exercise

1. Conduct comparative studies of empirical models of radio wave attenuation Okamura-Hata, COST 231-Hata and COST 231 Walvis-Ikegami for given characteristics of the communication channel for option 4 methodological instructions;

3. Prepare a work report with the following sections: 1) assignment, 2) theoretical part (text attached) and 3) research results - two figures with three graphs each.

Note: the calculation of the COST231Walfisch-Ikegami model is performed only for the case of line of sight.

Theoretical part

Basic provisions

Research on the propagation of radio waves in urban environments is of great importance in the theory and technology of communications. Indeed, the largest number of residents (potential subscribers) live in cities, and the conditions for propagation of radio waves differ significantly from propagation in free space and semi-free space. In the latter case, propagation over a regular earth's surface is understood when the radiation pattern does not intersect with the earth's surface. In this case, with directional antennas, the attenuation of radio waves is determined by the formula:

L = 32,45 + 20(lgd km + lgf MHz) – 10lgG per – 10lgG per, dB =

= L 0 - 10lgG per – 10lgG per, dB. (1)

Where L 0 – basic attenuation of free space, dB;

d km– distance between transmitter and receiver, km;

f MHz – operating frequency, MHz;

G lane And G pr are the gains of the transmitting and receiving antennas, respectively, dBi.

Major weakening L 0 is determined with isotropic antennas, which radiate uniformly in all directions and also receive. Therefore, attenuation occurs due to the dissipation of energy into space and a small supply to receiving antenna. When using directional antennas with their main beams oriented towards each other, the attenuation decreases in accordance with equation (1).

The objective of the study is to determine the radio channel carrying the message (radio signal), which provides the required quality and reliability of communication. The communication channel in urban conditions is not a deterministic quantity. Except direct channel There is interference between the transmitter and receiver due to numerous reflections from the ground, walls and roofs of buildings, as well as the passage of a radio signal through buildings. Depending on the relative position of the transmitter and receiver, there may be cases where there is no direct channel and behind received signal The receiver has to read the signal with the highest intensity. IN mobile communications, when the antenna of the subscriber receiver is at a height of 1 - 3 meters from the ground, these cases are dominant.

The statistical nature of received signals requires assumptions and constraints within which decisions can be made. The main assumption is the stationarity of the random process with the independence of interference noise from each other, that is, the absence of mutual correlation. The implementation of such requirements led to

division of urban radio communication channels into three main types: Gaussian, Ricean and Rayleigh channels.

The Gaussian channel is characterized by the presence of a dominant direct beam and low interference. The mathematical expectation of radio signal attenuation is described by the normal law. This channel is inherent television signals from a television tower when receiving on collective antennas on residential buildings. The Rice channel is characterized by the presence of direct rays, as well as reflected and transmitted rays through buildings and the presence of diffraction on buildings. The mathematical expectation of radio signal attenuation is described by the Rice distribution. This channel is inherent in networks with an elevated antenna above buildings in urban areas.

The Rayleigh channel is characterized by the absence of direct rays and the radio signal reaches the mobile station due to reflections. The mathematical expectation of the radio signal attenuation is described by the Rayleigh distribution. This channel is typical for cities with high-rise buildings.

Channel types and their density functions are taken into account when developing signal propagation models in urban environments. However, generalized statistics are not enough when calculating specific propagation conditions, under which signal attenuation depends on frequency, antenna height and building characteristics. Therefore, when implementing cellular communication and the need for frequency spatial planning, experimental studies of attenuation began to be conducted in various cities and propagation conditions. The first research results focused on mobile cellular communications appeared in 1989 (W.C.Y.Lee). However, even earlier, in 1968 (Y. Okumura) and in 1980 (M. Hata), the results of studies of the attenuation of radio waves in the city, focused on mobile trunking communications and television broadcasting, were published.

Further research was carried out with the support of the International Telecommunications Union (ITU) and was aimed at clarifying the conditions of applicability of the models.

Below we consider the models that have become most widespread in the design of communication networks for urban environments.

Units of measurement of radio signal levels

In practice, two types of units of measurement are used to evaluate the level of radio signals: 1) based on power units and 2) based on voltage units. Since the power at the output of the transmitter antenna is many orders of magnitude higher than the power at the input of the receiver antenna, multiple units of power and voltage are used.

Unit multiples are expressed in decibels (dB), which are relative units. Power is usually expressed in milliwatts or watts:

P dBm = 10 log (P/1 mW),(2)

P dBW = 10 log (P/ 1 W).(3)

For example, a power equal to 100 W in these units will be equal to: 50 dBm or 20 dBW.

Voltage units are based on 1 µV (microvolt):

U dBµV = 20 log (U/ 1 µV). (4)

For example, a voltage of 10 mV is equal to 80 dBµV in given relative units.

Relative power units are used, as a rule, to express the transmitter radio signal level, relative voltage units are used to express the receiver signal level. The relationship between the sizes of relative units can be obtained based on the equation P=U2/R or U 2 = PR, Where R is the input impedance of the antenna, matched with the line leading to the antenna. Taking the logarithm of the above equations, and taking into account equations (2) and (4), we obtain:

1 dBm = 1 dBµV – 107 dB at R= 50 Ohm; (5a)

1 dBm = 1 dBµV – 108.7 dB at R= 75Ohm. (5 B)

To express the transmitter power, the characteristic is often used - effective radiated power - ERP. This is the transmitter power taking into account the gain (GN = G) antennas:

ERP (dBW) = P (dBW) + G (dBi). (6)

For example, a 100 W transmitter drives an antenna with a gain of 12 dBi. Then EIM = 32 dBW, or 1.3 kW.

When calculating the coverage area of ​​a cellular base station or the coverage area of ​​an terrestrial television transmitter, one should take into account the antenna gain, that is, use the effective radiated power of the transmitter.

Antenna gain has two units of measurement: dBi (dBi)– gain relative to an isotropic antenna and dBd– gain relative to the dipole. They are related to each other by the relationship:

G (dBi) = G (dBd) + 2.15 dB. (7)

It should be taken into account that the antenna gain of the subscriber station is usually assumed to be zero.

Okamura-Hata model

The primary version of the model by Okamura and his co-authors is designed for the following application conditions: frequency range (150 - 1500) MHz, distance between mobile and base stations - from 1 to 100 km, base station antenna height - from 30 to 1000 m.

The model is based on comparing the attenuation in the city with the attenuation in free space, taking into account correction components depending on the frequency, height of the antennas of the base and mobile stations. The components are presented in the form of graphs. Long distances and heights of base stations are more suitable for television broadcasting than for cellular communications. In addition, the resolution of the graphs is low and less convenient than the analytical description.

Hata approximated Okamura's graphs with analytical relationships, reduced the frequency range to 1500 MHz (Okamura's was overestimated and did not meet the required reliability of the attenuation assessment), reduced the range of distances from one to twenty kilometers, and also reduced the height of the base station antenna to 200 meters and made clarifications into some components of Okamura's model. As a result of the Hata modernization, the model was named Okamura-Hata and is popular for assessing the attenuation of TV signals and in cellular communications in the range up to 1000 MHz.

For the city, power weakening L in decibels (dB) is described by the empirical formula:

L,dB=69.55 + 26.16 lgf - 13.83 lg +(44.9-6,55 lg d– a( ), (8)

Where f– frequency in MHz,

d- distance between the base and subscriber (mobile) stations in km,

The height of the antennas at the base and subscriber stations.

In formula (8) the component a() determines the influence of the height of the subscriber station antenna on the attenuation of signal power.

For an average city and an average building height, this component is determined by the formula:

a( ) = (1.1 lgf – 0.7)– 0.8, dB. (9)

For a city with high buildings a() is determined by the formula:

a( ) = 8,3 (log 1.54) 2 – 1.1 for f< 400 МГц; (10)

a( ) = 3,2 (lg 11.75) 2 – 5 for f> 400 MHz. (eleven)

In suburban areas, signal propagation losses depend more on frequency than on the height of the subscriber station antenna, and therefore, the component Δ is added to equation (8) taking into account equation (9) L,dB, defined by the equation:

Δ L,dB = - 5,4 – (lg (0.036 f)) 2. (12)

In open areas Δ L,dB for isotropic antennas is described by the equation:

Δ L,dB = - 41 – 4,8 (lgf) 2 + 18,33lgf. (13)

The disadvantage of the Okamura-Hata model is that the frequency range is limited to 1500 MHz and it cannot be used for distances less than one kilometer.

As part of the European Union's COST 231 project (Cooperation for Scientific and Technical Research), two models were developed that addressed the noted shortcomings of the Okamura-Hata model. These models are discussed below.

Model COST231-Hata

1 , < 200m, 1 < < 10m.